Device and Method for Measuring a Phase Deviation

ABSTRACT

A device for measuring a phase deviation has a sampler having a first input for a periodic measurement signal having a steady-state or a varying frequency, a second input for a reference signal replicating an idealized phase trajectory of the periodic measurement signal, and an output for a sample value of the reference signal sampled by use of the periodic measurement signal. A reference signal generator having an output coupled to the second input of the sampler is provided. Further, provision is made for a phase deviation identifier having an input coupled to the output of the sampler.

This application claims priority from German Patent Application No. 102006 031 351.8, which was filed on Jul. 6, 2006, and is incorporatedherein in its entirety by reference.

TECHNICAL FIELD

The present invention refers to a method and a device for measuring aphase deviation, or difference, in particular to a device for measuringa linearity of a frequency deflection, such as is used in automotiveradar systems, for example.

BACKGROUND

In automotive engineering, so-called FMCW radar systems (FMCW=frequencymodulated continuous wave) are used in driver assistance systems, forexample, to reduce the number of car accidents, among other things. In aFMCW radar system, a linearly frequency-modulated high-frequency signal(HF signal) is used. Thus, a time-dependent transmitting frequencyf_(HF)(t) of the HF signal linearly increases in a time interval Δt byan amount Δf_(HF), for example, or is deflected by Δf_(HF). Thisfrequency deflection is referred to as a so-called frequency sweep.Through a run time t_(d) during a signal propagation of the HF signal toa reflector, the transmitting frequency of the HF signal changes in themeantime, to f_(HF)(t+t_(d)) due to the frequency deflection, so thatone can obtain with a mixer a low-frequency signal with a frequencyf_(NF)(t+t_(d))=f_(HF)(t+t_(d))−f_(HF)(t) from the difference betweenthe current transmitting frequency f_(HF)(t+t_(d)) and the receivingfrequency f_(HF)(t) reflected by the reflector. The frequencyf_(NF)(t+t_(d)) is proportional to a reflector distance d. Thus, in FMCWradar systems, the run time t_(d) is converted to the frequencyf_(NF)(t+t_(d)). If the frequency sweep is linear, the frequencyf_(NF)(t+t_(d)) of the low-frequency mix signals will remain constantduring the sweep operation.

In FMCW radar systems, the linearity properties of the transmittedfrequency sweep are of great importance. Nowadays, typical frequencysweep bandwidths Δf_(HF) range from several hundred MHz to some GHz. Forautomotive radar applications, for example, a frequency band at 77 GHzis reserved. In comparison to a center frequency and the bandwidthΔf_(HF) of the frequency sweep, a non-linearity of the frequency sweepis very small and, for this reason, difficult to measure.

SUMMARY OF THE INVENTION

According to one embodiment, the present invention includes a device formeasuring a phase deviation with a sampler having a first input for aperiodic measurement signal comprising a steady-state or a variablefrequency, a second input for a reference signal replicating anidealized phase trajectory of the measurement signal, and an output fora sample value of the reference signal sampled by use of the measurementsignal, a reference signal generator with an output coupled to thesecond input of the sampler, and a phase deviation identifier with aninput coupled to the output of the sampler.

Embodiments of the present invention further provide a method formeasuring a phase deviation comprising a step of providing a periodicmeasurement signal comprising a steady-state or a variable frequency, astep of providing a reference signal replicating an idealized phasetrajectory of the measurement signal, a step of sampling the referencesignal by use of the measurement signal for generating a sample value ofthe reference signal, and a step of identifying the phase deviation ofthe measurement signal from the reference signal of the sample value ofthe reference signal.

BRIEF DESCRIPTION OF THE DRAWINGS

In the following, embodiments of the present invention will be explainedin detail with reference to the accompanying drawings, wherein:

FIG. 1 shows a conventional arrangement for a linearity measurement of afrequency sweep with a frequency converter and a digital storageoscilloscope;

FIG. 2 shows a diagrammatic block diagram of a conventional arrangementfor linearity measurement of a frequency sweep with a phase frequencydetector;

FIG. 3 shows a diagrammatic block diagram of a device for measuring aphase deviation according to an embodiment of the present invention;

FIG. 4 a shows a diagrammatic illustration of a frequency sweep plottedversus time;

FIG. 4 b shows a diagrammatic illustration of a measurement signal withvariable frequency;

FIG. 4 c shows a phase diagram for explaining of the measurementprinciple of the phase deviation according to an embodiment of thepresent invention; and

FIG. 5 shows a diagrammatic block diagram of a device for measuring aphase deviation according to an embodiment of the present invention withexternal wiring for a frequency conversion of the measurement signal.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

It should be noted that with respect to the following description,identical functional elements or functional elements operating in thesame way have identical reference numbers in the different embodiments,and, thus, the descriptions of these functional elements in thedifferent embodiments illustrated in the following are interchangeable.

Known systems for linearity measurement of a frequency sweep are basedon the use of so-called digital storage oscilloscopes (DSO). Such asystem is illustrated by way of example in FIG. 1.

FIG. 1 shows a signal source 100 coupled to a frequency divider 110 onthe output side. The output of the frequency divider 110 forms a firstinput of a mixer 120. An output of a local oscillator 130 forms a secondinput of the mixer 120. An output of the mixer 120 is coupled to alow-pass filter 140. An output of the low-pass filter 140 forms an inputof a digital storage oscilloscope 150 coupled to a controller means 160.

A simple and known approach to linearity measurement of a frequencysweep is to down-convert a time-dependent frequency f_(HF)(t) of theoutput signal s_(HF)(t) of the signal source 100 with the frequencydivider 110 by a factor N, for example. The measurement signaldown-converted in its frequency by the factor N and comprising an atleast approximately linear frequency sweep, i.e., a linear frequencydeflection, is further down-converted by means of the mixer 120 and thelocal oscillator 130 comprising a frequency f_(LO), and is sampled withthe digital storage oscilloscope 150 after low-pass filtration with thelow-pass filter 140. Thus, the mixed and low-pass filtered measurementsignal s_(meas)(t) comprises a frequency f_(meas)(t)=f_(HF)(t)/N−f_(LO).

A phase information may now be determined from the so-called analyticalsignal of the measurement signal s_(meas)(t), for example. One obtainsthe analytical signal by adding an imaginary portion resulting from theHilbert-transform of the real measurement signal s_(means)(t) to themeasurement signal s_(meas)(t). To calculate a phase error of theanalytical signal of the measurement signal, a mathematically generatedideal phase trajectory φ_(ref)(t) of the analytical signal, for example,is adapted to the measured data, and the different signal of the idealphase trajectory φ_(ref)(t) is compared with the phase trajectoryφ_(meas)(t) of the measurement signal.

Instead of the digital storage oscilloscope, an analog-to-digitalconverter, for example, may also be used. In the end, however, a complexand expensive signal management system still remains necessary to obtainthe result of the linearity measurement of the frequency sweep.

The use of a phase frequency detector is a further common approach tolinearity measurement of a frequency sweep. A diagrammatic block diagramof such a measurement system is shown in FIG. 2.

FIG. 2 shows a measurement signal source 100 coupled to a first input ofa phase frequency detector 200. A reference signal source 210 is coupledto a second input of the phase frequency detector 200. An output of thephase frequency detector 200 is wired to a digital storage oscilloscope,or an analog-to-digital converter, 150 controlled by a controller means160.

In the measurement system shown in FIG. 2, the reference signal source210 generates a reference signal s_(ref)(t) having an idealizedfrequency trajectory f_(ref)(t), or phase trajectory φ_(ref)(t), of themeasurement signal s_(meas)(t) generated by the measurement signalsource 100. The idealized frequency trajectory fret) means a requested,i.e. for example absolutely linear, frequency trajectory. On the basisof the measurement signal s_(meas)(t) and the reference signals_(ref)(t), the phase frequency detector 200 identifies a signal, suchas a current or a voltage, for example, proportional to a phasedifference φ_(diff)(t) of both of the signals, which is digitalized bymeans of the digital storage oscilloscope, or the analog-to-digitalconverter, 150, and is then further processed.

As in the measurement system described in the foregoing on the basis ofFIG. 1, this implementation has the disadvantage that the realization isvery expensive with respect to hardware and software.

Finally, the principle of FMCW radar systems itself may also be used forlinearity measurement of a frequency sweep. For this purpose, themeasurement signal s_(meas)(t) is mixed with the frequency rampresulting from the frequency deflection, such as a time-shifted versions_(meas)(t+t_(d)) thereof, for example. For this purpose, a coaxialcable may be used, for example, as a delay line with a known electricallength and a mixer mixing both of the time-shifted signals s_(meas)(t)and s_(meas)(t+t_(d)). In an ideal linear frequency sweep, thelow-frequency signal resulting at the mixer output comprises only asingle component at a frequency f_(NF)=f_(meas)(t+t_(d))−f_(meas)(t)corresponding to the time delay, whereas in the case of a non-linearityof the frequency sweep, the spectrum of the resulting signal isbroadened. Here, too, a sampling must be performed using ananalog-to-digital converter, and the result is to be evaluated by meansof software in a PC, for example.

After known systems for linearity measurement of a frequency sweep havebeen described in the foregoing on the basis of FIG. 1 and FIG. 2, theconcept for linearity measurement of a frequency sweep according to anembodiment of the present invention is to be illustrated in more detailon the basis of FIG. 3 to FIG. 5.

A measurement method according to one embodiment of the presentinvention is based on the principle of a direct digital synthesizer(DDS). A direct digital synthesizer numerically calculates in a clockcycle of the duration T_(clk) a phase φ of a 2π periodic signal using aso-called phase accumulator. A so-called tuning word forms a phaseincrement Δφ of the phase accumulator. For example, in a clock cycle n,the phase φ(n·T_(clk)) of the phase accumulator is increased by thephase increment Δφ, thus, φ(n·T_(clk))=φ((n−1)T_(clk))+Δφ. A digitalphase word of the phase accumulator consists of a specified number ofbits, such as j bits. Each time the phase accumulator overflows, i.e. ina transition from φ(n·T_(clk))=2^(j)−1 to φ((n+1)·T_(clk)), a completeperiod of the periodic signal is generated. For this reason, the phaseincrement Δφ of the phase accumulator and a clock frequencyf_(clk)=1/T_(clk) of the direct digital synthesizer define an outputfrequency f_(out) generated by the direct digital synthesizer. Byincreasing the tuning word, i.e., the phase increment Δφ, from one clockcycle to the next, a linear frequency sweep may be synthesized, forexample.

According to embodiments of the present invention, the output of thedigital phase accumulator is sampled at times of zero crossings of therising signal edge of the periodic measurement signal. In the process,the phase accumulator generates the reference frequency sweep by makingavailable phase values of the reference sweep at a high digitalresolution and with a clock rate f_(clk) suitable for the bandwidth ofthe frequency sweep. Since the phase of the measurement signal at thetime of the sampling comprises a value which is a multiple of 2π, thevalue of the phase accumulator at this sample time represents a measureof a phase deviation of the frequency sweep of the measurement signalfrom the ideal linear frequency sweep of the reference signal.

FIG. 3 shows a diagrammatic block diagram of a device for measuring aphase deviation according to an embodiment of the present invention.

Device 300 comprises a sampler 310 including a first input 310 a, asecond input 310 b, and an output 310 c. Device 300 further includes areference signal generator 320 with an output 320 a coupled to thesecond input 310 b of the sampler 310. Device 300 further includes aphase deviation identifier 330 having an input 330 a coupled to theoutput 310 c of the sampler 310. As indicated by the dotted line 340,the phase deviation identifier may be coupled, e.g., to the referencesignal generator 320.

Via the input 310 a, a periodic measurement signal s_(meas)(t) that maycomprise a steady-state or a variable frequency f_(meas)(t) is suppliedto the sampler 310. A digital reference signal s_(ref)(t) generated bythe reference signal generator 320 with a clock frequency f_(clk) andreplicating an idealized frequency trajectory f_(ref)(t), or phasetrajectory φ_(ref)(t), of the analog periodic measurement signals_(meas)(t) present at the input 310 a is present at the second input310 b of the sampler 310, i.e. s_(ref)(t)=φ_(ref)(t).

One frequency trajectory f_(meas)(t), which is possible in principle, ofthe periodic measurement signal s_(meas)(t) present at the input 310 aof the sampler 310 is shown in FIG. 4 a. The graph marked with referencenumber 400 means an idealized frequency trajectory f_(ref)(t) of themeasurement signal s_(meas)(t), whereas the graph marked with referencenumber 410 represents a possible real frequency trajectory f_(meas)(t)of the measurement signal s_(meas)(t). On the basis of FIG. 4 a itshould be appreciated that in a linear frequency sweep, a frequency isincreased from a frequency f₁ to a frequency f₂ within a periodΔt=(T₂−T₁).

For better illustration, this connection is represented yet again inFIG. 4 b. FIG. 4 b shows a measurement, or an oscillation, signals_(meas)(t) whose oscillation frequency continuously increases.

A periodic signal of the form as is shown in FIG. 4 b, for example, ispresent as the measurement signal s_(meas)(t) at the input 310 a of thesampler 310 of the device 300. The times when the sampler 310 samplesthe reference signal s_(ref)(t) present at the input 310 b are marked,by way of example, with t₁ to t₄ in FIG. 4 b. According to oneembodiment of the present invention, the sampler 310 samples thereference signal s_(ref)(t) within a predetermined range of zerocrossings of the rising signal edge of the periodic measurement signals_(meas)(t) at the input 310 a, as is shown in FIG. 4 b. Thepredetermined range means that sampling is performed exactly at the zerocrossing of s_(meas)(t), for example, or within a range in which themagnitude |s_(meas)(t)| of the measurement signal comprises a valuesmaller than 10% of the amplitude of s_(meas)(t). Thus, the followingconditions are at least approximately satisfied at the sample times:

s _(meas)(t)=0 and  (1)

ds _(meas)(t)/dt>0  (2)

If conditions (1) and (2) are satisfied, the phase φ_(meas)(t) of theperiodic measurement signal s_(meas)(t) comprises a value, which atleast approximately corresponds to a multiple of 2π, i.e.φ_(meas)(t)=i*2π(i=0, 1, 2 . . . ). The measurement signal s_(meas)(t)will typically comprise a frequency response which does not take anideal linear course, as is indicated in FIG. 4 a by the graph withreference number 410.

By the reference signal generator 320, a digital reference signals_(ref)(t) replicating an idealized linear phase trajectory φ_(ref)(t)of the measurement signal is generated, as is indicated by referencenumber 400 in FIG. 4 a. This reference signal s_(ref)(t) is now sampledby the sampler 310 at those times, for example, when the measurementsignal s_(meas)(t) comprises a zero crossing of the rising signal edge,such as has been described in the foregoing and is indicated in FIG. 4b. Since the sample times comprise a generally non-constant timeinterval of 1/f_(meas)(t), the phase of the measurement signalφ_(meas)(t) is a multiple of 2π, the sampled phase value φ_(ref)(t) ofthe reference signal at these sample times represents a measure of aphase deviation of the measurement signal from the ideal referencefrequency sweep. For a better illustration, this connection is depictedin FIG. 4 c.

FIG. 4 c shows a phase diagram comprising a phase indicator 420comprising a position corresponding to a phase value according to amultiple of 2π. The phase indicator 420 corresponds to the phaseindicator of the measurement signal s_(meas)(t) at the sample times.FIG. 4 c further shows a phase indicator 430 a which deviates from thephase indicator 420 by a phase difference Δφ_(,) and a phase indicator430 b which deviates from the phase indicator 420 by a phase differenceΔφ₂. The positions of the phase indicators 430 a,b correspond topossible phase values of the reference signal φ_(ref)(t) at the sampletimes.

It is to be understood that in embodiments of the present invention,sample times other than the times of zero crossings of the rising signaledge are also possible. For example, the sampler 310 could sample thereference signal s_(ref)(t) within a predetermined range of zerocrossings of the falling signal edge of the periodic measurement signals_(meas)(t). The phase φ_(meas)(t) of the periodic measurement signals_(meas)(t) would then comprise a value which at least approximatelycorresponds to an odd-number multiple of π, i.e.φ_(meas)(t)=(2i+1)*π(i=0, 1, 2, . . . ). The sampler 310 could furthersample the reference signal s_(ref)(t) generally within a predeterminedrange of zero crossings of the periodic measurement signal s_(meas)(t).The phase φ_(meas)(t) of the periodic measurement signal s_(meas)(t)would then comprise a value which at least approximately corresponds toa multiple of π, i.e. φ_(meas)(t)=i*π(i=0, 1, 2, 3, . . . ).

With the phase deviation identifier 330 shown in FIG. 3, the phasedeviation between the measurement signal s_(meas)(t) and the referencesignal s_(ref)(t) may be determined as follows. The phase φ_(meas)(t) ofthe measurement signal s_(meas)(t) comprises, at the sample times, avalue which corresponds to a multiple of 2π, as is indicated in FIG. 4 cby the phase indicator 420. The phase value φ_(ref)(t) of the referencesignal of the phase accumulator at the sample times, or at the zerocrossings of the rising signal edge of the measurement signals_(meas)(t), represents the sampled measurement value

φ_(samp)(t)=φ_(ref)(t)−φ_(meas)(t),  (3).

the phase value φ_(meas)(t), related to a period of the measurementsignal, comprising at least approximately a value of zero due to thezero crossings, i.e. φ_(meas)(t)=0. The phase deviation φ_(diff)(t) maythus be determined according to

φ_(diff)(t)=φ_(meas)(t)−φ_(ref)(t)=−φ_(samp)(t)=−φ_(ref)(t)  (4).

As already described in the foregoing, this value φ_(diff)(t) is sampledat a sample frequency corresponding only to the momentary frequencyf_(meas)(t) of the measurement signal s_(meas)(t). By contrast, in theconcepts described on the basis of FIGS. 1 and 2 a sample frequencycorresponding to at least twice the frequency f_(meas)(t) of themeasurement signal s_(meas)(t) is used.

For most practical applications, this measurement data φ_(diff)(t),which is not uniformly sampled, has to be resampled again at a constantsample rate. However, with suitable signal processing in the phasedeviation identifier 330, it is also possible to work with themeasurement data φ_(diff)(t), which is non-uniformly sampled. Afrequency difference φ_(diff)(t) between the reference signal and themeasurement signal s_(meas)(t) may be determined by a numericdifferentiation of the phase measurement data, for example,φ_(diff)(t)=dφ_(diff)(t)/dt.

Embodiments of the present invention may allow a real-time measurementwith minimum hardware costs, and thus allow an in-system frequency sweepevaluation and possibly even a sweep linearization.

FIG. 5 shows a diagrammatic block diagram of a device for measuring aphase deviation according to a further embodiment of the presentinvention comprising external wiring to down-convert a frequency of ameasurement signal.

FIG. 5 shows a measurement signal source 100 connected to a frequencydivider 110. An output of the frequency divider 110 forms a first inputof a mixer 120, and an output of a stable local oscillator 130 forms asecond input of the mixer 120. An output of the mixer 120 forms an input310 a of a sampler 310 of the device 300. An output 320 a of a referencesignal generator 320 is coupled to a second input 310 b of the sampler310. The reference signal generator 320 comprises a phase incrementgenerator 500 coupled to a phase accumulator 510. The output of thephase accumulator forms the output of the reference signal generator320. The phase increment generator 500 is controlled by a controller330. An input of the controller 330 is coupled to an output 310 c of thesampler 310.

The time-varying frequency f_(HF)(t) of the periodic measurement signals_(HF)(t) with the sweep bandwidth Δf_(HF) of the measurement signalsource 100 is down-converted, by means of the frequency divider 110 andthe mixer 120, to a frequency f_(meas)(t) and a bandwidth Δf_(meas) thatare suitable for conventional digital logic technologies. Further, themeasurement signal s_(meas)(t) is down-converted in its frequency isused as a clock signal for the sampler 310, as described in theforegoing, to sample the momentary values φ_(ref)(t) of the phaseaccumulator 510 and then to determine therefrom a phase deviation of themeasurement signal from the reference signal in accordance with aprocedure according to an embodiment of the present invention. The phaseincrement generator 500 and the phase accumulator 510 generate thefrequency sweep ideally expected by the measurement signal s_(meas)(t).The measurement signal source 100 may be the transmitter of anautomotive radar system, for example. Automotive radar systems work in afrequency band at 77 GHz, for example, and generate frequency sweeps ofa bandwidth Δf_(HF) of approximately 1 GHz. Since a frequency f_(ref)generated by a direct digital synthesizer is typically within a range ofseveral hundred MHz to 1 GHz, the original measurement signal s_(HF)(t)of the measurement signal source 100 is down-converted in its frequency.The clock rate f_(clk) of the phase accumulator 510 should be largeenough to cover the necessary signal bandwidth of the down-convertedmeasurement signal s_(meas)(t). If the output of the phase accumulator510 comprises a word width of j bits, a time-varying frequency of thereference signal may be represented by means of a time-varying phaseincrement Δφ_(ref)(t) according to

${f_{ref}(t)} = \frac{\Delta \; {{\phi_{ref}(t)} \cdot f_{clk}}}{2^{j}}$

In this context, the time response of the phase increment Δφ_(ref)(t) ofthe phase increment generator 500 is controlled by the controller 330,for example.

Thus, embodiments of the present invention have the advantage thatin-system measurements of a linearity of a frequency sweep over largebandwidths are allowed with few hardware requirements. Thus, embodimentsof the present invention may allow real-time measurement with minimumhardware expenditure, and thus, an in-system frequency sweep evaluationand possibly even a sweep linearization.

In particular, it should be understood that depending on thecircumstances, the inventive scheme may also be implemented in tosoftware. The implementation may be made on a digital storage medium, inparticular a disc or a CD with control signals that can be read outelectronically and can co-operate with a programmable computer system sothe respective method is carried out. In general, the invention alsoexists as a computer program product comprising a program code, storedon a machine-readable carrier, to perform the inventive method, when runon a computer. In other words, the invention may be realized as acomputer program comprising a program code for performing the method,when the run on a computer.

While this invention has been described in terms of several embodiments,there are alterations, permutations, and equivalents which fall withinthe scope of this invention. It should also be noted that there are manyalternative ways of implementing the methods and compositions of thepresent invention. It is therefore intended that the following appendedclaims be interpreted as including all such alterations, permutations,and equivalents as fall within the true spirit and scope of the presentinvention.

1. A device for measuring a phase deviation, the device comprising: asampler comprising a first input for a periodic measurement signalcomprising a steady-state or a varying frequency, a second input for areference signal replicating an idealized phase trajectory of theperiodic measurement signal, and an output for a sample value of thereference signal sampled by use of the periodic measurement signal; areference signal generator comprising an output coupled to the secondinput of the sampler; and a phase deviation identifier comprising aninput coupled to the output of the sampler.
 2. The device according toclaim 1, wherein the periodic measurement signal present at the firstinput of the sampler comprises a linear frequency sweep, or a linearlyincreasing or falling sweep frequency.
 3. The device according to claim1, wherein the sampler can sample the reference signal within apredetermined range of a zero crossing of a rising or falling signaledge of the periodic measurement signal.
 4. The device according toclaim 1, wherein the reference signal generator comprises a phaseincrement generator comprising a phase increment output, and a phaseaccumulator comprising an input coupled to the phase increment output,and a reference signal output coupled to the second input of thesampler.
 5. The device according to claim 1, wherein a clock frequencyof the reference signal generator is larger than a highest frequency ofthe periodic measurement signal.
 6. The device according to claim 1,wherein the phase deviation identifier identifies a phase deviationφ_(diff)(t) of a phase φ_(meas)(t) of the periodic measurement signalfrom a phase φ_(ref)(t) of the reference signal according to:φ_(diff)(t)=φ_(meas)(t)−φ_(ref)(t), t corresponding to sample times ofthe sampler.
 7. A device for measuring a phase deviation of a phase of aperiodic measurement signal from a phase of a reference signalreplicating an idealized phase trajectory of the periodic measurementsignal with a linearly varying sweep frequency, the device comprising: asampler comprising a first input for the periodic measurement signal, asecond input for the reference signal, and an output for a sample valueof the reference signal, the sample value of the reference signal beingsampled within a predetermined range of a zero crossing of a rising orfalling signal edge of the periodic measurement signal; a phaseincrement generator comprising a phase increment output; a phaseaccumulator comprising a phase increment input coupled to the phaseincrement output of the phase increment generator, and comprising areference signal output coupled to the second input of the sampler; anda phase deviation identifier comprising an input coupled to the outputof the sampler.
 8. The device according to claim 7, wherein a clockfrequency of the phase accumulator is larger than a highest frequency ofthe periodic measurement signal.
 9. The device according to claim 7,wherein the phase deviation identifier identifies a phase deviationφ_(diff)(t) of a phase φ_(meas)(t) of the periodic measurement signalfrom a phase φ_(ref)(t) of the reference signal according to:φ_(diff)(t)=φ_(meas)(t)−φ_(ref)(t), t corresponding to sample times ofthe sampler.
 10. A device for measuring a phase deviation, comprising:means for providing a periodic measurement signal comprising asteady-state or a varying frequency; means for providing a referencesignal replicating an idealized phase trajectory of the periodicmeasurement signal; means for sampling the reference signal by use ofthe periodic measurement signal for generating a sample value of thereference signal; and means for identifying, from the sample value ofthe reference signal, the phase deviation of the periodic measurementsignal from the reference signal.
 11. The device according to claim 10,wherein the periodic measurement signal provided by the means forproviding comprises a linear frequency sweep.
 12. The device accordingto claim 10, wherein the means for sampling samples the reference signalwithin a predetermined range of a zero crossing of a rising or fallingsignal edge of the periodic measurement signal.
 13. The device accordingto claim 10, wherein the means for providing a reference signal furthercomprises a means for providing a phase increment and a means forproviding an accumulated phase.
 14. The device according to claim 10,wherein a clock frequency of the means for providing the referencesignal is larger than the highest frequency of the periodic measurementsignal.
 15. The device according to claim 10, wherein the means foridentifying the phase deviation identifies the phase deviationφ_(diff)(t) of a phase φ_(meas)(t) of the periodic measurement signalfrom a phase φ_(ref)(t) of the reference signal according to:φ_(diff)(t)=φ_(meas)(t)−φ_(pref)(t), t corresponding to sample times ofthe sampler.
 16. A method for measuring a phase deviation, the methodcomprising: providing a periodic measurement signal comprising asteady-state or varying frequency; providing a reference signalreplicating an idealized phase trajectory of the periodic measurementsignal; sampling the reference signal using the periodic measurementsignal to generate a sample value of the reference signal; andidentifying, from the sample value of the reference signal, the phasedeviation of the periodic measurement signal from the reference signal.17. The method according to claim 16, wherein providing the periodicmeasurement signal is performed such that the periodic measurementsignal comprises a linear frequency sweep, or a linearly increasing orfalling sweep frequency.
 18. The method according to claim 16, whereinsampling of the reference signal is performed within a predeterminedrange of a zero crossing of a rising or falling signal edge of theperiodic measurement signal.
 19. The method according to claim 16,wherein providing the reference signal further comprises providing aphase increment and providing an accumulated phase.
 20. The methodaccording to claim 16, wherein providing the reference signal furthercomprises providing a clock frequency, the clock frequency being largerthan a highest frequency of the periodic measurement signal.
 21. Themethod according to claim 16, wherein identifying the phase deviation ofthe periodic measurement signal is performed such that the phasedeviation φ_(diff)(t) of a phase φ_(meas)(t) of the periodic measurementsignal from a phase φ_(ref)(t) of the reference signal is identifiedaccording to:φ_(diff)(t)=φ_(meas)(t)−φ_(ref)(t), t corresponding to sample times ofthe sampler.
 22. A computer program comprising a program code forperforming a method for measuring a phase deviation, the methodcomprising: providing a periodic measurement signal comprising asteady-state or varying frequency; providing a reference signalreplicating an idealized phase trajectory of the periodic measurementsignal; sampling the reference signal by use of the periodic measurementsignal for generating a sample value of the reference signal; andidentifying, from the sample value of the reference signal, the phasedeviation of the periodic measurement signal from the reference signal,when the computer program runs on a computer.